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19-3289; Rev 0; 8/04
KIT ATION EVALU E AILABL AV
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
General Description Features
3V to 28V Supply Voltage Range 1.2% Accurate Over Temperature Adjustable Output Voltage Down to 0.6V 200kHz to 500kHz Switching Frequency Adjustable Temperature-Compensated Hiccup Current Limit Lossless Peak Current Sensing Monotonic Startup into Prebias Output (MAX8576/MAX8578) Startup Overvoltage Protection (MAX8577/MAX8579) Enable/Shutdown Adjustable Soft-Start
MAX8576-MAX8579
The MAX8576-MAX8579 synchronous PWM buck controllers use a hysteretic voltage-mode control algorithm to achieve a fast transient response without requiring loop compensation. The MAX8576/MAX8577 contain an internal LDO regulator allowing the controllers to function from only one 3V to 28V input supply. The MAX8578/MAX8579 do not contain the internal LDO and require a separate supply to power the IC when the input supply is higher than 5.5V. The MAX8576- MAX8579 output voltages are adjustable from 0.6V to 0.9 x VIN at loads up to 15A. Nominal switching frequency is programmable over the 200kHz to 500kHz range. High-side MOSFET sensing is used for adjustable hiccup current-limit and short-circuit protection. The MAX8576/MAX8578 can start up into a precharged output without pulling the output voltage down. The MAX8577/MAX8579 have startup output overvoltage protection (OVP), and will pull down a precharged output.
Applications
Motherboard Power Supplies AGP and PCI-Express Power Supplies Graphic-Card Power Supplies Set-Top Boxes Point-of-Load Power Supplies
Ordering Information
PART MAX8576EUB MAX8577EUB MAX8578EUB MAX8579EUB TEMP RANGE -40C to +85C -40C to +85C -40C to +85C -40C to +85C PIN-PACKAGE 10 MAX(R) 10 MAX 10 MAX 10 MAX
Typical Operating Circuit
INPUT UP TO 28V
FB OCSET SS IN
VL
MAX8576 MAX8577
DH OUTPUT 0.6V TO 0.9 x VIN LX
GND
DL
BST
MAX is a registered trademark of Maxim Integrated Products, Inc.
Pin Configurations appear at end of data sheet. 1
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
ABSOLUTE MAXIMUM RATINGS
IN to GND (MAX8576/MAX8577) ...........................-0.3V to +30V VL to GND (MAX8576/MAX8577).............................-0.3V to +6V IN to VL (MAX8576/MAX8577) ...............................-0.3V to +30V VCC to GND (MAX8578/MAX8579) ..........................-0.3V to +6V SS to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V SS to GND (MAX8578/MAX8579)...............-0.3V to (VCC + 0.3)V DL to GND (MAX8576/MAX8577) ...............-0.3V to (VVL + 0.3)V DL to GND (MAX8578/MAX8579) ..............-0.3V to (VCC + 0.3)V BST to GND ............................................................-0.3V to +36V BST to LX..................................................................-0.3V to +6V LX to GND .....................-1V (-2.5V for <50ns Transient) to +30V DH to LX..................................................-0.3V to +(VBST + 0.3)V FB to GND ................................................................-0.3V to +6V EN to GND (MAX8578/MAX8679EUB) .....................-0.3V to +6V OCSET to GND (MAX8576/MAX8677) ........-0.3V to (VIN + 0.3)V OCSET to GND (MAX8578/MAX8679) ...................-0.3V to +30V OCSET to LX (MAX8576/MAX8677) ............-0.6V to (VIN + 0.3)V OCSET to LX (MAX8578/MAX8679) .......................-0.6V to +30V DH and DL Continuous Current ............................250mA RMS Continuous Power Dissipation (TA = +70C) 10-Pin MAX (derate 5.6mW/C above +70C) ...........444mW Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) ................................+300C
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = 12V (MAX8576/MAX8577 only), 4.7F capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01F capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER SUPPLY VOLTAGES IN Supply Voltage VCC Input Voltage VL Output Voltage VL Maximum Output Current VL or VCC Undervoltage Lockout (UVLO) MAX8576/MAX8577 IN = VL (MAX8576/MAX8577) MAX8576/MAX8577 IVL = 10mA (MAX8576/MAX8577) MAX8576/MAX8577 Rising Falling Hysteresis No switching, VFB = 0.65V (MAX8576/MAX8577) Supply Current VEN = 0V or VFB = 0.65V, no switching (MAX8578/MAX8579) REGULATOR Output Regulation Accuracy Output Regulation Hysteresis FB Propagation Delay Overvoltage-Protection (OVP) Threshold High-Side Current-Sense Program Current (Note 2) TA = +85C TA = +25C 42.5 VFB peak (Note 1) FB falling to DL falling FB rising to DH falling 0.70 0.593 12.5 0.6 20 50 70 0.75 60 50 57.5 0.80 0.607 28.0 V mV ns V A VIN = 12V VIN = VVL = 5V VIN = VVL = 3.3V VCC = 5V VCC = 3.3V 5.5 3.0 3.0 4.75 20 2.75 2.4 2.8 2.45 350 0.6 1.1 0.6 0.6 0.6 2 3 2 2 2 mA 2.90 2.5 5.0 28.0 5.5 5.5 5.25 V V V mA V mV CONDITIONS MIN TYP MAX UNITS
2
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V (MAX8576/MAX8577 only), 4.7F capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01F capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = 0C to +85C, unless otherwise noted. Typical values are at TA = +25C.)
PARAMETER High-Side Current-Sense Overcurrent Trip Adjustment Range Soft-Start Internal Resistance Fault Hiccup Internal SS Pulldown VLX < VOCSET and VFB < VSS Current DRIVER SPECIFICATIONS DH Driver Resistance DL Driver Resistance Dead Time DH Minimum On-Time DL Minimum On-Time BST Current EN Input Voltage Low Input Voltage High THERMAL SHUTDOWN Thermal Shutdown Rising temperature, hysteresis = 20C (typ) +160 C VCC = 3V (MAX8578/MAX8579) VCC = 5.5V (MAX8578/MAX8579) 1.5 0.7 V V Normal operation Current fault VBST - VLX = 5.5V, VLX = 28V, VFB < VSS Sourcing current Sinking current Sourcing current Sinking current DH low to DL high and DL low to DH high (adaptive) 2.6 1.9 2.6 1.1 40 140 120 580 1.65 245 220 4.0 3.0 4.0 2.0 ns ns ns mA VIN - VOCSET CONDITIONS MIN 0.05 45 80 250 TYP MAX 0.40 125 UNITS V k nA
MAX8576-MAX8579
ELECTRICAL CHARACTERISTICS
(VIN = 12V (MAX8576/MAX8577 only), 4.7F capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01F capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = -40C to +85C, unless otherwise noted. Note 3)
PARAMETER SUPPLY VOLTAGES IN Supply Voltage VCC Input Voltage VL Output Voltage VL Maximum Output Current MAX8576/MAX8577 IN = VL, MAX8576/MAX8577 MAX8576/MAX8577 IVL = 10mA, MAX8576/MAX8577 MAX8576/MAX8577 5.5 3.0 3.0 4.75 20 28.0 5.5 5.5 5.25 V V V mA CONDITIONS MIN TYP MAX UNITS
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3
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V (MAX8576/MAX8577 only), 4.7F capacitor from VL (MAX8576/MAX8577 only) or VCC (MAX8578/MAX8579 only) to GND; VCC = VEN = 5V (MAX8578/MAX8579 only); 0.01F capacitor from SS to GND; VFB = 0.65V; VBST = 5V; VLX = VGND = 0V; VOCSET = 11.5V; DH = unconnected; DL = unconnected; TA = -40C to +85C, unless otherwise noted. Note 3)
PARAMETER VL or VCC Undervoltage Lockout (UVLO) Rising Falling No switching, VFB = 0.65V (MAX8576/MAX8577) Supply Current VEN = 0V or VFB = 0.65V, no switching (MAX8578/MAX8579) REGULATOR Output Regulation Accuracy Overvoltage-Protection (OVP) Threshold High-Side Current-Sense OverCurrent Trip Adjustment Range DRIVER SPECIFICATIONS DH Driver Resistance DL Driver Resistance DH Minimum On-Time DL Minimum On-Time EN Input Voltage Low Input Voltage High VCC = 3V, MAX8578/MAX8579 VCC = 5.5V, MAX8578/MAX8579 1.5 0.7 V V Normal operation Sourcing current Sinking current Sourcing current Sinking current 4 3.0 4.0 2.0 245 220 ns ns VIN - VOCSET VFB peak 0.591 0.70 0.05 0.607 0.80 0.40 V V V VIN = 12V VIN = VVL = 5V VIN = VVL = 3.3V VCC = 5V VCC = 3.3V CONDITIONS MIN 2.75 2.40 TYP MAX 2.90 2.55 2 3.5 2 2 2 mA UNITS V
Note 1: Guaranteed by design. Note 2: This current linearly compensates for the MOSFET temperature coefficient. Note 3: Specifications to -40C are guaranteed by design and not production tested.
4
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Typical Operating Characteristics
(TA = +25C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 2)
MAX8576-79 toc01
EFFICIENCY vs. LOAD CURRENT (CIRCUIT OF FIGURE 3)
MAX8576-79 toc02
LOAD REGULATION (CIRCUIT OF FIGURE 2)
1.83 1.82 OUTPUT VOLTAGE (V) 1.81 1.80 1.79 1.78 1.77
MAX8576-79 toc03
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.1 1 10 LOAD CURRENT (A) VOUT = 1.5V VOUT = 3.3V
VOUT = 2.5V
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 VOUT = 1.5V VOUT = 2.5V VOUT = 3.3V VOUT = 1.8V
1.84
VOUT = 1.8V
VIN = 12V 100
VIN = 12V 0.1 1 LOAD CURRENT (A) 10
1.76 1.75 0 5 10 15 LOAD CURRENT (A)
0
LINE REGULATION (CIRCUIT OF FIGURE 2)
MAX8576-79 toc04
LINE REGULATION (CIRCUIT OF FIGURE 3)
MAX8576-79 toc05
SWITCHING FREQUENCY vs. INPUT VOLTAGE (CIRCUIT OF FIGURE 3)
550 SWITCHING FREQUENCY (kHz) 500 450 400 350 300 250 200
MAX8576-79 toc06
1.85 1.84 OUTPUT VOLTAGE (V) 1.83 1.82 1.81 15A LOAD 1.80 1.79 5 10 15 INPUT VOLTAGE (V) 20 0A LOAD
1.86 1.84 OUTPUT VOLTAGE (V) 1.82 1.80 1.78 1.76 1.74 1.72 1.70 5A LOAD NO LOAD
600
25
5
10
15 INPUT VOLTAGE (V)
20
25
5
10
15 INPUT VOLTAGE (V)
20
25
LOAD TRANSIENT (CIRCUIT OF FIGURE 2)
MAX8576-79 toc07
LOAD TRANSIENT (CIRCUIT OF FIGURE 3)
MAX8576-79 toc08
5A IOUT IOUT 12A 6A 2.5A
VOUT
50mV/div AC-COUPLED
VOUT
50mV/div AC-COUPLED
40s/div
40s/div
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5
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Typical Operating Characteristics (continued)
( TA = +25C, unless otherwise noted.)
POWER-UP VIN (CIRCUIT OF FIGURE 3)
MAX8576-79 toc09
POWER-UP VCC (CIRCUIT OF FIGURE 3)
MAX8576-79 toc10
VIN
10V/div 0
VIN
10V/div 0
VCC
5V/div 1V/div 5A/div 0
VCC
5V/div 1V/div
VOUT ILX
VOUT 5A/div ILX 0
400s/div
400s/div
POWER-DOWN VCC (CIRCUIT OF FIGURE 3)
MAX8576-79 toc11
POWER-UP (CIRCUIT OF FIGURE 2)
MAX8576-79 toc12
VIN
10V/div 0
VIN
10V/div 0
VCC VOUT
5V/div VOUT 1V/div 1V/div 0 10A/div ILX 0
ILX
5A/div 0
400s/div
1ms/div
POWER-DOWN (CIRCUIT OF FIGURE 2, MAX8576)
MAX8576-79 toc13
STARTUP AND SHUTDOWN (CIRCUIT OF FIGURE 3)
MAX8576-79 toc14
VIN
5V/div
VEN
2V/div 0
0 VOUT 1V/div 0 ILX 10A/div 0
VDL VOUT
10V/div 0 1V/div 0
ILX
5A/div 0
4ms/div
400s/div
6
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Typical Operating Characteristics (continued)
( TA = +25C, unless otherwise noted.)
STARTUP AND SHUTDOWN (CIRCUIT OF FIGURE 2)
MAX8576-79 toc15
MONOTONIC OUTPUT-VOLTAGE RISE (CIRCUIT OF FIGURE 2, MAX8576)
MAX8576-79 toc16
VGS(Q3)
10V/div 0 0.5V/div 0 2V/div 0 10A/div 0
VIN VOUT 1.5V VLX
10V/div 0.5V/div 20V/div
VSS VOUT
ILX VDL 40ms/div 1ms/div 5V/div 0
NONMONOTONIC OUTPUT-VOLTAGE RISE (CIRCUIT OF FIGURE 2, MAX8577)
MAX8576-79 toc17
SHORT CIRCUIT AND RECOVERY (CIRCUIT OF FIGURE 2)
MAX8576-79 toc18
VIN VOUT 1.5V VLX
10V/div 0.5V/div 20V/div
VIN
10V/div
IIN IOUT
2A/div
10A/div 2V/div 0
VOUT VDL 1ms/div 5V/div 0 10ms/div
OUTPUT OVERVOLTAGE PROTECTION (CIRCUIT OF FIGURE 2)
MAX8576-79 toc19
VDH
20V/div 0 5V/div 0 1V/div 0
VDL
VOUT
VFB
0.5V/div 0 200s/div
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7
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Pin Description
PIN MAX8576/ MAX8577 1 MAX8578/ MAX8579 1 NAME FUNCTION
FB
Feedback Input. Regulates at VFB = 0.59V. Connect FB to a resistor-divider to set the output voltage. See the Setting the Output Voltage section. Soft-Start. Use an external capacitor (CSS) to adjust the soft-start time. An internal 80k resistor gives approximately 4ms soft-start time for a 0.01F external capacitor. An internal 250nA current sink in hiccup mode gives approximately 10% duty cycle during fault conditions. Internal 5V Linear-Regulator Output. Bypass with a 4.7F or larger ceramic capacitor. Must be connected to IN for operation from a 3.3V to 5.5V input. Supply Input (3V to 5.5V). Bypass with a 4.7F or larger ceramic capacitor to GND. Ground Low-Side Gate-Drive Output. Drives the synchronous-rectifier MOSFET. Boost-Capacitor Connection for High-Side Gate-Drive Output. Connect a 0.1F ceramic capacitor from BST to LX and a Schottky or switching diode and a 4.7 series resistor from BST to VL (MAX8576/MAX8577) or VCC (MAX8578/MAX8579). See Figure 4. External Inductor Connection. Connect LX to the junctions of the MOSFETs and inductor. High-Side Gate-Drive Output. Drives the high-side MOSFET. Supply Voltage Input of the Internal Linear Regulator (3V to 28V). Connect to VL for operation from 3V to 5.5V input. Connect a 0.47F or larger ceramic capacitor from IN to GND. Enable Input. A logic low on EN shuts down the converter and discharges the soft-start capacitor. Drive high or connect to IN for normal operation. Overcurrent-Limit Set. Programs the high-side peak current-limit threshold by setting the maximum-allowed VDS voltage drop across the high-side MOSFET. Connect a resistor from IN to OCSET; an internal 50A current sink sets the maximum voltage drop relative to VIN. See the Setting the Current Limit section.
2
2
SS
3 -- 4 5
-- 3 4 5
VL VCC GND DL
6
6
BST
7 8 9 --
7 8 -- 9
LX DH IN EN
10
10
OCSET
8
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
BST FAULT
OCSET LX
MAX8578/ MAX8579 EN
GND
0.3 DHI DH FB DRIVERS OVP 0.75V LOGIC LX
SS
SS RAMP
DLI DL
0.05V IN MAX8576 MAX8577 VL VL REG VLOK POK
MAX8576-MAX8579
VCC MAX8578 MAX8579
REF
REFOK
GND
Figure 1. Functional Diagram
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9
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
INPUT 9V TO 24V C1 C11 C2 R1 1 C4 2 OFF ON R4 Q3 R5 C6 4 GND 3 VL SS IN 9 C5 FB OCSET 10
R2
C3
C8
C12
MAX8576 MAX8577
DH
8
Q1 R3 L1
R7
OUTPUT 1.8V/12A
LX
7 C7 C9 C13
Q2
C10
5
DL
BST
6
R6 R8
D1
12V INPUT, 1.8V/12A OUTPUT (fS = 300kHz) CIRCUIT IS TARGETED FOR 10.8V TO 13.2V INPUT. HOWEVER, INPUT RANGE OF 9V TO 24V IS POSSIBLE FOR IC EVALUATION. 30V RATED MOSFET MUST BE INSTALLED IF INPUT IS RAISED ABOVE 16V. ALL OTHER COMPONENTS CAN REMAIN UNCHANGED.
Figure 2. MAX8576/MAX8577 Typical Application Circuit
C21 ON OFF R9 1 C16 2 SS EN 9 FB OCSET 10
R10
C14
C15
C19
3 C17 4
VCC
MAX8578 MAX8579
DH
8
Q4 R11 L2 VOUT 1.8V/5A C20 C23
Q5
GND
LX
7 C18 C22
5
DL
BST
6
R12 R13
D2
12V INPUT, 1.8V/5A OUTPUT (fS = 500kHz, ALL CERAMIC) CIRCUIT IS TARGETED FOR 10.8V TO 13.2V INPUT. HOWEVER, INPUT RANGE OF 9V TO 24V IS POSSIBLE FOR IC EVALUATION. 30V RATED MOSFET MUST BE INSTALLED IF INPUT IS RAISED ABOVE 16V. ALL OTHER COMPONENTS CAN REMAIN UNCHANGED.
Figure 3. MAX8578/MAX8579 Typical Application Circuit
10
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
MAX8576/MAX8577 External Component List
COMPONENTS QTY DESCRIPTION/VENDOR PART NUMBER 470F, 35V aluminum electrolytic capacitors Sanyo 35MV470WX 10F, 25V X7R ceramic capacitor 0.01F, 10V X7R ceramic capacitor 1F, 35V X7R ceramic capacitor 4.7F, 6.3V X5R ceramic capacitor 0.1F, 10V X7R ceramic capacitors 0.027F, 25V X7R ceramic capacitor 2200F, 6.3V aluminum electrolytic capacitors Rubycon 6.3MBZ2200M10X20 0.01F, 25V X5R ceramic capacitor 3300pF, 6.3V X5R ceramic capacitor High-speed diode, 100V, 250mA Philips BAS316 (SOD-323) 1.8H, 14A, 3.48m Panasonic ETQP2H1R8BFA 30V, 12.5m (max), SO-8 International Rectifier IRF7821 30V, 3.7m, SO-8 International Rectifier IRF7832 2N7002 SOT-23 6.04k 1% resistor 5.11 k 1% resistor 12.4k 1% resistor 1k 5% resistor 20k 5% resistor 2 5% resistor 10 5% resistor 4.7 5% resistor C23 D2 L2 Q4 Q5 R9 R10 R11 R12 R13 1 1 1 1 1 1 1 1 1 1 C21 C22 1 0 C20 1 COMPONENT C14 C15 C16 C17 C18 C19
MAX8578/MAX8579 External Component List
QTY 1 1 1 1 1 1 DESCRIPTION/VENDOR PART NUMBER 10F, 25V X5R ceramic capacitor 1F, 25V X5R ceramic capacitor 4700pF, 10V X7R ceramic capacitor 4.7F, 6.3V X5R ceramic capacitor 0.1F, 10V X7R ceramic capacitor 0.01F, 25V X7R ceramic capacitor 47F, 6.3V, ESR = 5m, ceramic capacitor Taiyo Yuden JMK432476MM 0.01F, 25V X5R ceramic capacitor Optional (47F, 6.3V, ESR = 5m ceramic capacitor Taiyo Yuden JMK432476MM) 1000pF, 25V X5R ceramic capacitor High-speed diode, 100V, 250mA Philips BAS316 (SOD-323) 2.2H, 7.3A, 9.8m Sumida CDEP104L-2R2 30V, 18m (max), SO-8 International Rectifier IRF7807Z 30V, 9.5m, SO-8 International Rectifier IRF7821 6.04k 1% resistor 2.49k 1% resistor 12.4k 1% resistor 2 5% resistor 4.7 5% resistor
MAX8576-MAX8579
C1, C2 C3 C4 C5 C6 C7, C12 C8 C9, C10 C11 C13 D1 L1 Q1 Q2 Q3 R1 R2 R3 R4 R5 R6 R7 R8
2 1 1 1 1 2 1 2 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1
Detailed Description
The MAX8576-MAX8579 synchronous PWM buck controllers use Maxim's proprietary hysteretic voltagemode control algorithm to achieve fast transient response without any loop-compensation requirement. The controller drives a pair of external n-channel power MOSFETs to improve efficiency and cost. The
MAX8576/MAX8577 contain an internal linear lowdropout (LDO) regulator allowing the controller to operate from a single 3V to 28V input supply. The MAX8578/MAX8579 do not contain the internal LDO and require a separate supply to power the IC when the input supply is higher than 5.5V. The MAX8576- MAX8579 output voltages are adjustable from 0.6V to 0.9 x VIN at loads up to 15A.
11
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Nominal switching frequency is programmable over the 200kHz to 500kHz range. High-side MOSFET sensing is used for adjustable hiccup current-limit and short-circuit protection. The MAX8576/MAX8578 can start up into a precharged output without pulling the output voltage down. The MAX8577/MAX8579 have startup output overvoltage protection (OVP). The MAX8578/MAX8579 have a logic-enable input to turn on and off the output. The MAX8576/MAX8577 are turned off by pulling SS low with an external small n-channel MOSFET (see Figure 2). supply from 3V to 5.5V connected to VCC if the input voltage is greater than 5.5V.
Undervoltage Lockout
If VL (MAX8576/MAX8577) or VCC (MAX8578/MAX8579) drops below 2.45V (typ), the MAX8576-MAX8579 assume that the supply voltage is too low for proper circuit operation, so the UVLO circuitry inhibits switching and forces the DL and DH gate drivers low for the MAX8576/MAX8578, and DH low and DL high for the MAX8577/MAX8579. After VIN rises above 2.8V (typ), the controller goes into the startup sequence and resumes normal operation.
DC-DC Converter Control Architecture
A proprietary hysteretic-PWM control scheme ensures high efficiency, fast switching, and fast transient response. This control scheme is simple: when the output voltage falls below the regulation threshold, the error comparator begins a switching cycle by turning on the high-side switch. This switch remains on until the minimum on-time expires and the output voltage is in regulation or the current-limit threshold is exceeded. Once off, the high-side switch remains off until the minimum off-time expires and the output voltage falls below the regulation threshold. During this period, the lowside synchronous rectifier turns on and remains on until the voltage at FB drops below its regulation threshold. The internal synchronous rectifier eliminates the need for an external Schottky diode.
Output Overvoltage Protection
The MAX8576-MAX8579 output overvoltage protection is provided by a glitch-resistant comparator on FB with a trip threshold of 750mV (typ). The overvoltage-protection circuit is latched by an OVP fault, terminating the run cycle and setting DH low and DL high. The fault is cleared by toggling EN or UVLO. Output OVP is active whenever the internal reference is in regulation.
Startup and Soft-Start
The soft-start sequence is initiated upon initial powerup, recovering from UVLO, or driving EN (MAX8578/ MAX8579) high from a low state, or releasing SS (MAX8576/MAX8577) from a low state. The external soft-start capacitor (CSS) is connected to an internal resistor-divider that exponentially charges the capacitor to 0.6V, with an SS ramp interval of 5 x RC or 4ms per 0.01F. SS is one input to the internal voltage error comparator, while FB is the other input. The output voltage fed back to FB tracks the rising SS voltage. Switching commences immediately if VFB is initially less than VSS; if VFB is greater than VSS, DH remains low until V FB is less than V SS . DL remains low in the MAX8576/MAX8578. This prevents the converter from operating in reverse. However, DL is high before startup in the MAX8577/MAX8579 to enable OVP protection in case the high-side MOSFET is shorted.
Voltage-Positioning Load Regulation
As seen in Figures 2 and 3, the MAX8576-MAX8579 use a unique feedback network. By taking feedback from the LX node through R3 (R11 for the MAX8578/MAX8579), the usual phase lag due to the output capacitor does not exist, making the loop stable for either electrolytic or ceramic output capacitors. This configuration causes the output voltage to shift by the inductor DC resistance multiplied by the load current. This voltage-positioning load regulation greatly reduces overshoot during load transients, which effectively halves the peak-to-peak outputvoltage excursions compared to traditional step-down converters. See the Load Transient graphs in the Typical Operating Characteristics.
Enable
Connecting EN to GND or logic low places the MAX8578/MAX8579 in shutdown mode. In shutdown, DH and DL are forced low, and the voltage at SS is discharged with a 250nA current, resulting in a rampdown interval of approximately 10x the soft-start ramp-up interval. VSS must fall to within 50mV of GND before another cycle can commence. SS (MAX8576/ MAX8577) or EN (MAX8578/MAX8579) do not need to be cycled after an overcurrent event. Connect EN to IN or logic high for normal operation. To shut down the MAX8576/MAX8577, use an external circuit connected
Internal 5V Linear Regulator
All MAX8576/MAX8577 functions are powered from the on-chip, low-dropout 5V regulator with the input connected to IN. Bypass the regulator's output (VL) with a 1F or greater ceramic capacitor. The capacitor must have an equivalent series resistance (ESR) of no greater than 10m. When VIN is less than 5.5V, short VL to IN. The MAX8578/MAX8579 do not have the onchip 5V regulator and must use a separate external
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
to SS. See Figure 2 for details. The maximum on-resistance of the small external n-channel MOSFET should be less than 40 so that the SS voltage is below 10mV.
Current-Limit Circuit
Current limit is set externally with a resistor from OCSET to the drain of the high-side n-channel MOSFET that is normally connected to the input supply. The resistor programs the high-side peak current limit by setting the maximum-allowed V DS(ON) voltage drop across the high-side MOSFET. An internal 50A current sink sets the maximum voltage drop relative to V IN . If V FB < 300mV, any overcurrent event (VDS of the high-side n-channel MOSFET is larger than the limit programmed at OCSET) immediately sets DH low and terminates the run cycle. If VFB > 300mV and an overcurrent event is detected, DH is immediately set low and four sequential overcurrent events terminate the run cycle. Once the run cycle is terminated, the SS capacitor is slowly discharged through the internal 250nA current sink to provide a hiccup current-limit effect. Choosing the proper value resistor is discussed in the Setting the Current Limit section.
MAX8576-MAX8579
Synchronous-Rectifier Driver (DL)
Synchronous rectification reduces conduction losses in the rectifier by replacing the normal Schottky catch diode with a low-resistance MOSFET switch. The MAX8576-MAX8579 also use the synchronous rectifier to ensure proper startup of the boost gate-driver circuit. The DL low-side waveform is always the complement of the DH high-side drive waveform (with controlled dead time to prevent cross-conduction or shoot-through). A dead-time circuit monitors the DL output and prevents the high-side MOSFET from turning on until DL is fully off. For the dead-time circuit to work properly, there must be a low-resistance, low-inductance path from the DL driver to the MOSFET gate. Otherwise, the sense circuitry in the MAX8576-MAX8579 may interpret the MOSFET gate as off when gate charge actually remains. Use very short, wide traces (50 mils to 100 mils wide if the MOSFET is 1in from the device). The dead time at the other edge (DH turning off) is also determined through gate sensing.
Switching Frequency
Nominal switching frequency is programmable over the 200kHz to 500kHz range. This allows tradeoffs in efficiency, switching frequency, inductor value, and component size. Faster switching frequency allows for smaller inductor values but does result in some efficiency loss. Inductor-value calculations are provided in the Inductor Value section. The switching frequency is tuned by the selection of the feed-forward capacitor (CFF). See the Feed-Forward Capacitor section.
High-Side Gate-Drive Supply (BST)
Gate-drive voltage for the high-side n-channel switch is generated by a flying-capacitor boost circuit (Figure 4). The capacitor between BST and LX is charged from the IN supply up to VIN minus the diode drop while the lowside MOSFET is on. When the low-side MOSFET is switched off, the stored voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage (VGS) for the high-side MOSFET. The controller then closes an internal switch between BST and DH to turn the high-side MOSFET on.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation in the MAX8576-MAX8579. When the junction temperature exceeds T J = +160C, an internal thermal sensor shuts down the IC, allowing the IC to cool. The thermal sensor turns the IC on again after the junction temperature cools to +140C, resulting in a pulsed output during continuous thermal-overload conditions.
Design Procedures
IN
Setting the Output Voltage
BST DH N
MAX8576- MAX8579
LX
DL
N
Select an output voltage between 0.6V and 0.9 x VIN by connecting FB to a resistive voltage-divider between LX and GND (see Figures 2 and 3). Choose R1 for approximately 50A to 150A bias current in the resistive divider. A wide range of resistor values is acceptable, but a good starting point is to choose R1 as 6.04k. Then, R3 is given by:
Figure 4. DH Boost Circuit ______________________________________________________________________________________ 13
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
VOUT + 0.01V + (RDC x 0.5 x IOUTMAX ) R3 = R1 x - 1 VFB
where VFB = 0.590V, RDC is the DC resistance of the output inductor, IOUTMAX is the maximum output current. The term 0.01V is to reflect 1/2 of the feedbackthreshold hysteresis.
junction temperature, typically +85C to +125C depending on the application. ROCSET is calculated using the VDS(ON)MAX with the following formula: VDS(ON)MAX ROCSET = 50A A 0.01F ceramic capacitor is required in parallel with ROCSET to decouple high-frequency noise.
Inductor Value
The inductor value is bounded by two operating parameters: the switching frequency and the inductor peakto-peak ripple current. The peak-to-peak ripple current is typically in the range of 20% to 40% of the maximum output current. The equation below defines the inductance value: VOUT x (VIN - VOUT ) L= VIN x fS x ILOAD(MAX ) x LIR where LIR is the ratio of inductor current ripple to DC load current and fS is the switching frequency. A good compromise between size, efficiency, and cost is an LIR of 30%. The selected inductor must have a saturated current rating above the sum of the maximum output current and half of the peak-to-peak ripple current. The DC current rating of the inductor must be above the maximum output current to keep the temperature rise within the desired range. In addition, the DC resistance of the inductor must meet the requirement below: RDC VOUT IOUTMAX
MOSFET Selection
The MAX8576-MAX8579 drive two external, logic-level, n-channel MOSFETs as the circuit switching elements. The key selection parameters are: 1) On-resistance (RDS(ON)): the lower, the better. 2) Maximum drain-to-source voltage (V DSS): should be at least 20% higher than the input supply rail at the high-side MOSFET's drain. 3) Gate charges (Qg, Qgd, Qgs): the lower, the better. For a 3.3V input application, choose a MOSFET with a rated RDS(ON) at VGS = 2.5V. For a 5V input application, choose the MOSFETs with rated RDS(ON) at VGS 4.5V. For a good compromise between efficiency and cost, choose the high-side MOSFET (N1) that has conduction losses equal to switching loss at nominal input voltage and output current. The selected high-side MOSFET (N1) must have RDS(ON) that satisfies the current-limit-setting condition above. For N2, make sure that it does not spuriously turn on due to dV/dt caused by N1 turning on as this results in shoot-through current degrading the efficiency. MOSFETs with a lower Qgd / Qgs ratio have higher immunity to dV/dt. For proper thermal-management design, the power dissipation must be calculated at the desired maximum operating junction temperature, maximum output current, and worst-case input voltage (for the low-side MOSFET, worst case is at VIN(MAX); for the high-side MOSFET, it could be either at VIN(MAX) or VIN(MIN)). N1 and N2 have different loss components due to the circuit operation. N2 operates as a zero-voltage switch; therefore, major losses are: the channel-conduction loss (P N2CC ) and the body-diode conduction loss (PN2DC). V PN2CC = 1 - OUT x ILOAD2 x RDS(ON) VIN Use RDS(ON) at TJ(MAX). PN2DC = 2 x ILOAD x VF x t dt x fS
where VOUT is the maximum-allowed output-voltage drop from no load to full load (IOUTMAX).
Setting the Current Limit
Resistor R2 (R7 for the MAX8577/MAX8579) of Figure 2 (Figure 3 for the MAX8577/MAX8579) sets the current limit and is connected between OCSET and the drain of the high-side n-channel MOSFET. An internal 50A current sink sets the maximum voltage drop across the high-side n-channel MOSFET relative to VIN. The maximum VDS drop needs to be determined. This is calculated by: VDS(ON)MAX = IDS(MAX) x RDS(ON)MAX IDS(MAX) must be equal or greater than the maximum peak inductor current at the maximum output current. Use RDS(ON)MAX at the highest expected operating
14
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
where VF is the body-diode forward-voltage drop, tDT is the dead time between N1 and N2 switching transitions (40ns typ), and fS is the switching frequency. N1 operates as a duty-cycle control switch and has the following major losses: the channel-conduction loss (PN1CC), the VL overlapping switching loss (PN1SW), and the drive loss (PN1DR). N1 does not have bodydiode conduction loss because the diode never conducts current. V PN1CC = OUT x ILOAD2 x RDS(ON) VIN Use RDS(ON) at TJ(MAX). Qgs + Qgd PN1SW = VIN x ILOAD x x fS IGATE where IGATE is the average DH driver output-current capability determined by: IGATE 0.5 x VL RDH + RGATE
Input Capacitor
The input filter capacitor reduces peak currents drawn from the power source and reduces noise and voltage ripple on the input caused by the circuit's switching. The input capacitor must meet the ripple-current requirement (IRMS) imposed by the switching currents defined by the following equation: IRMS = ILOAD x VOUT x (VIN - VOUT ) VIN
MAX8576-MAX8579
I RMS has a maximum value when the input voltage equals twice the output voltage (VIN = 2 x VOUT), so IRMS(MAX) = ILOAD / 2. Ceramic capacitors are recommended due to their low ESR and ESL at high frequency, with relatively lower cost. Choose a capacitor that exhibits less than 10C temperature rise at the maximum operating RMS current for optimum long-term reliability.
Output Capacitor
The key selection parameters for the output capacitor are the actual capacitance value, the ESR, the equivalent series inductance (ESL), and the voltage-rating requirements. These parameters affect the overall stability, output voltage ripple, and transient response. The output ripple has three components: variations in the charge stored in the output capacitor, the voltage drop across the capacitor's ESR, and the ESL caused by the current into and out of the capacitor. The maximum output ripple voltage can be estimated by: VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL) The output voltage ripple as a consequence of the ESR and output capacitance is: VRIPPLE(ESR) = IP-P x ESR VRIPPLE(C) = IP-P COUT x fS
where RDH is the high-side MOSFET driver's on-resistance (2 typ) and RGATE is the internal gate resistance of the MOSFET (approximately 2). RGATE PN1DR = Qg x VGS x fS x RGATE + RDH where VGS is approximately equal to VL. In addition to the losses above, allow about 20% more for additional losses due to MOSFET output capacitances and N2 body-diode reverse-recovery charge dissipated in N1 that exists, but is not well defined in the MOSFET data sheet. Refer to the MOSFET data sheet for thermal-resistance specification to calculate the PC board area needed to maintain the desired maximum operating junction temperature with the above calculated power dissipations. To reduce EMI caused by switching noise, add 0.1F ceramic capacitor from the high-side switch drain to the low-side switch source or add resistors in series with DH and DL to slow down the switching transitions. However, adding series resistors increases the power dissipation of the MOSFET, so be sure this does not overheat the MOSFET. The minimum load current must exceed the high-side MOSFET's maximum leakage current over temperature if fault conditions are expected.
V VRIPPLE(ESL) = IN x ESL L V - VOUT VOUT IP-P = IN x V fS x L IN where IP-P is the peak-to-peak inductor current (see the Inductor Value section). These equations are suitable for initial capacitor selection, but final values should be
15
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
chosen based on a prototype or evaluation circuit. As a general rule, a smaller current ripple results in less output voltage ripple. Since the inductor ripple current is a factor of the inductor value and input voltage, the output voltage ripple decreases with larger inductance and increases with higher input voltages. For reliable and safe operation, ensure that the capacitor's voltage and ripple-current ratings exceed the calculated values. The response of the MAX8576-MAX8579 to a load 1transient depends on the selected output capacitors. After a load transient, the output voltage instantly changes by ESR times ILOAD. Before the controller can respond, the output voltage deviates further depending on the inductor and output capacitor values. The controller responds immediately as the output voltage deviates from its regulation limit (see the Typical Operating Characteristics). The MAX8576-MAX8579 are compatible with both aluminum electrolytic and ceramic output capacitors. Due to the limited capacitance of a ceramic capacitor, it is typically used for a higher switching frequency and lower output current. Aluminum electrolytic is more applicable to frequencies up to 300KHz and can support higher output current with its much higher capacitance value. Due to the much higher ESL and ESR of the aluminum electrolytic capacitor, an RC filter (R7 and C12 of Figure 2) is required to prevent excessive ESL and ESR ripple from tripping the feedback threshold prematurely.
MAX8576-MAX8579
The resistor for critical dampening (RSNUB) is equal to 2 x fR x LPAR. Adjust the resistor value up or down to tailor the desired damping and the peak voltage excursion. The capacitor (CSNUB) should be at least 2 to 4 times the value of CPAR to be effective. The power loss of the snubber circuit is dissipated in the resistor (PRSNUB) and can be calculated as: PRSNUB = CSNUB x (VIN )2 x fSW where VIN is the input voltage and fSW is the switching frequency. Choose an RSNUB power rating that meets the specific application's derating rule for the power dissipation calculated.
Feed-Forward Capacitor
The feed-forward capacitor, C8 (Figure 2, MAX8576/ MAX8577 with aluminum electrolytic output capacitor), or C19 (Figure 3, MAX8578/MAX8579 with ceramic output capacitor), dominantly affects the switching frequency. Choose a ceramic X7R capacitor with a value given by: C8 = or C19 = V 1 1 V x - 120ns x IN x 39.5 x 1- OUT RFB FS VOUT VIN 1 V 1 V x - 120ns x IN x 49.5 x 1- OUT RFB FS VOUT VIN
MOSFET Snubber Circuit
Fast-switching transitions cause ringing because of resonating circuit parasitic inductance and capacitance at the switching nodes. This high-frequency ringing occurs at LX's rising and falling transitions and can interfere with circuit performance and generate EMI. To dampen this ringing, a series RC snubber circuit is added across each switch. Below is the procedure for selecting the value of the series RC circuit: 1) Connect a scope probe to measure V LX to GND, and observe the ringing frequency, fR. 2) Find the capacitor value (connected from LX to GND) that reduces the ringing frequency by half. The circuit parasitic (CPAR) at LX is then equal to 1/3 the value of the added capacitance above. The circuit parasitic inductance (LPAR) is calculated by: LPAR = 1 (2fR )2 x CPAR
where FS is the desired switching frequency, and RFB is the parallel combination of the two feedback dividerresistors (R1 and R3 of Figure 2, and R9 and R11 of Figure 3). Select the closest standard value to C8 and C19 as possible. The output inductor and output capacitor also affect the switching frequency, but to a much lesser extent. The equations for C8 and C19 above should yield within 30% of the desired switching frequency for most applications. The values of C8 and C19 can be increased to lower the frequency, or decreased to raise the frequency for better accuracy.
Application Information
PC Board Layout Guidelines
Careful PC board layout is critical to achieve low switching losses and clean, stable operation. The switching power stage requires particular attention. Follow these guidelines for good PC board layout:
16
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3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers
1) Place IC decoupling capacitors as close to IC pins as possible. Place the input ceramic decoupling capacitor directly across and as close as possible to the high-side MOSFET's drain and the low-side MOSFET's source. This is to help contain the high switching current within this small loop. 2) For output current > 10A, a four-layer PC board is recommended. Pour a ground plane in the second layer underneath the IC to minimize noise coupling. 3) Input, output, and VL capacitors are connected to the power ground plane with the exception of C12 and C22. These capacitors and all other capacitors are connected to the analog ground plane. 4) Make the connection from the current-limit setting resistor directly to the high-side MOSFET's drain to minimize the effect of PC board trace resistance and inductance. 5) Place the MOSFET as close as possible to the IC to minimize trace inductance. If parallel MOSFETs are used, keep the gate connection to both gates equal. 6) Connect the drain leads of the power MOSFET to a large copper area to help cool the device. Refer to the power MOSFET data sheet for the recommended copper area. 7) Place the feedback components as close to the IC pins as possible. The feedback divider-resistor from FB to the output inductor should be connected directly to the inductor and not sharing with other connections to this node. 8) Refer to the EV kit for further guidelines.
MAX8576-MAX8579
Suggested External Component Manufacturers
MANUFACTURER Central Semiconductor Panasonic Sumida International Rectifier Kemet Taiyo Yuden TDK Rubycon COMPONENT Diodes Inductors Inductors MOSFETs Capacitors Capacitors Capacitors Capacitors WEBSITE www.centralsemi.com www.panasonic.com www.sumida.com www.irf.com www.kemet.com www.t-yuden.com www.component.tdk.com www.rubycon.com PHONE 631-435-1110 402-564-3131 847-956-0666 800-341-0392 864-963-6300 408-573-4150 888-835-6646 408-467-3864
Pin Configurations
TOP VIEW
FB 1 SS VL GND DL 2 3 4 5 10 OCSET 9 IN DH LX BST FB 1 SS VCC GND DL 2 3 4 5 10 OCSET 9 EN DH LX BST
MAX8576 MAX8577
8 7 6
MAX8578 MAX8579
8 7 6
MAX
MAX
Chip Information
TRANSISTOR COUNT: 2087 PROCESSS: BICMOS
______________________________________________________________________________________ 17
3V to 28V Input, Low-Cost, Hysteretic Synchronous Step-Down Controllers MAX8576-MAX8579
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information, go to www.maxim-ic.com/packages.)
e
10
4X S
10
INCHES MAX DIM MIN 0.043 A 0.006 A1 0.002 A2 0.030 0.037 D1 0.120 0.116 0.118 D2 0.114 E1 0.116 0.120 0.118 E2 0.114 0.199 H 0.187 L 0.0157 0.0275 L1 0.037 REF b 0.007 0.0106 e 0.0197 BSC c 0.0035 0.0078 0.0196 REF S 0 6
MILLIMETERS MAX MIN 1.10 0.15 0.05 0.75 0.95 3.05 2.95 2.89 3.00 3.05 2.95 2.89 3.00 4.75 5.05 0.40 0.70 0.940 REF 0.177 0.270 0.500 BSC 0.090 0.200 0.498 REF 0 6
H 0 0.500.1 0.60.1
1
1
0.60.1
TOP VIEW
BOTTOM VIEW
D2 GAGE PLANE A2 A b A1 D1
E2
c
E1 L1
L
FRONT VIEW
SIDE VIEW
PROPRIETARY INFORMATION TITLE:
PACKAGE OUTLINE, 10L uMAX/uSOP
APPROVAL DOCUMENT CONTROL NO. REV.
21-0061
I
1 1
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 (c) 2004 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.
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